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1 ltc1701 typical applicatio n u descriptio u applicatio s u 1mhz step-down dc/dc converter in sot-23 n pdas/palmtop pcs n digital cameras n cellular phones n portable media players n pc cards n handheld equipment , ltc and lt are registered trademarks of linear technology corporation. n tiny 5-lead sot-23 package n uses tiny capacitors and inductor n high frequency operation: 1mhz n high output current: 500ma n low r ds(on) internal switch: 0.28 w n high efficiency: up to 94% n current mode operation for excellent line and load transient response n short-circuit protected n low quiescent current: 135 m a n low dropout operation: 100% duty cycle n ultralow shutdown current: i q < 1 m a n peak inductor current independent of inductor value n output voltages from 5v down to 1.25v the ltc ? 1701 is the industrys first 5-lead sot-23 step- down, current mode, dc/dc converter. intended for low to medium power applications, it operates from 2.5v to 5.5v input voltage range and switches at 1mhz, allowing the use of tiny, low cost capacitors and inductors 2mm or less in height. the output voltage is adjustable from 1.25v to 5v. a built-in 0.28 w switch allows up to 0.5a of output current at high efficiency. opti-loop tm compensation allows the transient response to be optimized over a wide range of loads and output capacitors. the ltc1701 incorporates automatic power saving burst mode tm operation to reduce gate charge losses when the load current drops below the level required for continuous operation. with no load, the converter draws only 135 m a. in shutdown, it draws less than 1 m a, making it ideal for current sensitive applications. in dropout, the internal p-channel mosfet switch is turned on continuously, thereby maximizing battery life. its small size and switching frequency enables the com- plete dc/dc converter function to consume less than 0.3 square inches of pc board area. v in i th /run sw v fb gnd ltc1701 d1 l1 4.7 h r4 1m c1: taiyo yuden jmk316bj106ml c2: sanyo poscap 6tpa47m d1: mbrm120l l1: sumida cd43-4r7 r3 5.1k c3 330pf c1 10 f + r2 121k r1 121k c2 47 f 1701 f01 v out (2.5v/ 500ma) v in 2.5v to 5.5v + figure 1. 2.5v/500ma step-down regulator efficiency curve burst mode and opti-loop are trademarks of linear technology corporation. load current (ma) 1 10 100 1000 efficiency (%) 100 95 90 85 80 75 70 1701 f01a v in = 3.3v features
2 ltc1701 order part number s5 part marking t jmax = 125 c, q ja = 250 c/w consult factory for industrial and military grade parts. ltkg ltc1701es5 absolute axi u rati gs w ww u package/order i for atio uu w (voltages referred to gnd pin) v in voltage (pin 5).......................................C 0.3v to 6v i th /run voltage (pin 4) ..............................C 0.3v to 3v v fb voltage (pin 3) ......................................C 0.3v to 3v v in C sw (max switch voltage) ................8.5v to C 0.3v operating temperature range (note 2) .. C 40 c to 85 c junction temperature (note 5) ............................. 125 c storage temperature range ................. C 65 c to 150 c lead temperature (soldering, 10 sec).................. 300 c (note 1) the l denotes the specifications which apply over the full operating temperature range, otherwise specifications are at t a = 25 c. v in = 3.3v, r ith/run = 1meg (from v in to i th /run) unless otherwise specified. (note 2) symbol parameter conditions min typ max units v in operating voltage range 2.5 5.5 v i fb feedback pin input current (note 3) 0.1 m a v fb feedback voltage (note 3) l 1.22 1.25 1.28 v d v line reg reference voltage line regulation v in = 2.5v to 5v (note 3) 0.04 0.1 %/v d v load reg output voltage load regulation measured in servo loop, v ith = 1.5v, (note 3) 0.01 0.70 % measured in servo loop, v ith = 1.9v, (note 3) C 0.80 C1.50 % input dc supply current (note 4) active mode v fb = 0v 185 300 m a sleep mode v fb = 1.4v 135 200 m a shutdown v ith/run = 0v 0.25 1 m a v ith/run run threshold high i th/run ramping down 1.4 1.6 v run threshold low i th/run ramping up 0.3 0.6 v i ith/run run pullup current v ith/run = 1v 50 100 300 m a i sw(peak) peak switch current threshold v fb = 0v 0.9 1.1 a r ds(on) switch on resistance v in = 5v, v fb = 0v 0.28 w v in = 3.3v, v fb = 0v 0.30 w v in = 2.5v, v fb = 0v 0.35 w i sw(lkg) switch leakage current v in = 5v, v ith/run = 0v, v fb = 0v 0.01 1 m a t off switch off-time 400 500 600 ns electrical characteristics note 1: absolute maximum ratings are those values beyond which the life of a device may be impaired. note 2: the ltc1701e is guaranteed to meet performance specifications from 0 c to 70 c. specifications over the C 40 c to 85 c operating temperature range are assured by design, characterization and correlation with statistical process controls. note 3: the ltc1701 is tested in a feedback loop which servos v fb to the midpoint for the error amplifier without r ith/run = 1mhz (v ith = 1.7v unless otherwise specified). note 4: dynamic supply current is higher due to the internal gate charge being delivered at the switching frequency. note 5: t j is calculated from the ambient t a and power dissipation p d according to the following formula: ltc1701es5: t j = t a + (p d ? 250 c/w) sw 1 gnd 2 v fb 3 5 v in 4 i th /run top view s5 package 5-lead plastic sot-23 see the application information section
3 ltc1701 load current (ma) 1 efficiency (%) 100 95 90 85 80 75 70 10 100 1000 1701 ?g01 v in = 5.0v v in = 3.3v v out = 2.5v figure 1 input voltage (v) 2 efficiency (%) 6 1701 ?g02 3 4 5 100 95 90 85 80 75 70 65 60 i load =100ma i load =10ma i load =1ma v out = 2.5v figure 1 input voltage (v) supply current ( m a) 1701 ?g03 300 250 200 150 100 50 0 2 3 4 5 6 active sleep supply voltage (v) on resistance (m w ) 1701 ?g04 370 350 330 310 290 270 250 2 3 4 5 6 load current (ma) 0 v out error (%) 200 400 1701 ?g05 0.60 0.40 0.20 0.00 ?.20 ?.40 ?.60 ?.80 ?.00 ?.20 ?.40 600 v out = 5.0v v out = 3.3v v in (v) v out error (%) 1701 ?g06 0.30 0.25 0.20 0.15 0.10 0.05 0 2 3 4 5 6 i load = 200ma i load = 400ma v in (v) 3.0 v out (v) 3.8 701 ?g07 3.2 3.4 3.6 3.4 3.3 3.2 3.1 3.0 2.9 2.8 2.7 2.6 i load = 100ma i load = 500ma i load = 200ma v out = 3.3v figure 1 1701 g09 v out 50mv/div ac coupled v in = 3.3v, v out = 2.5v circuit of figure 1 i load = 100ma to 500ma step i l 200ma/div typical perfor m a n ce characteristics uw efficiency vs load current efficiency vs input voltage dc supply current switch resistance vs supply voltage load regulation line regulation dropout characteristics start-up transient response 1701 g08 v out 1v/div v in = 3.3v, v out = 2.5v circuit of figure 1 r load = 6 w i th 2v/div i l 500ma/div
4 ltc1701 sw (pin 1): the switch node connection to the inductor. this pin swings from v in to a schottky diode (external) voltage drop below ground. the cathode of the schottky diode must be closely connected to this pin. gnd (pin 2): ground pin. connect to the (C) terminal of c out , the schottky diode and (C) terminal of c in . v fb (pin 3): receives the feedback voltage from the external resistive divider across the output. nominal volt- age for this pin is 1.25v. i th /run (pin 4): combination of error amplifier compen- sation point and run control input. the current compara- tor threshold increases with this control voltage. nominal voltage range for this pin is 1.25v to 2.25v. forcing this pin below 0.8v causes the device to be shut down. in shutdown all functions are disabled. v in (pin 5): main supply pin and the (+) input to the current comparator. must be closely decoupled to ground. pi fu ctio s uuu block diagra w + 1.25v bandgap reference i th /ref clamp shdn 50 a v ref v ref 1.4v 1.5v v ref (1.25v) + over voltage comp + i th comp + error amp (1.25v to 2.25v) + current comp v ref + current sense amp sw gnd 1701 bd v in off-timer and gate control logic gate driver v fb i th /run pulse stretcher v fb < 0.6v v in v in pin limit table nominal (v) absolute max (v) pin name description min typ max min max 1 sw switch node C 0.3 v in v in C 8.5 v in + 0.3 2 gnd ground pin 0 3v fb output feedback pin 0 1.25 1.35 C 0.3 3 4i th /run error amplifier compensation and run pin 0 2.25 C 0.3 3 5v in main power supply 2.5 5.5 C 0.3 6
5 ltc1701 operatio u the ltc1701 uses a contant off-time, current mode archi- tecture. the operating frequency is then determined by the off-time and the difference between v in and v out . to optimize efficiency, the ltc1701 automatically switches between continuous and burst mode operation. the output voltage is set by an external divider returned to the v fb pin. an error amplfier compares the divided output voltage with a reference voltage of 1.25v and adjusts the peak inductor current accordingly. main control loop during normal operation, the internal pmos switch is turned on when the v fb voltage is below the reference voltage. the current into the inductor and the load in- creases until the current limit is reached. the switch turns off and energy stored in the inductor flows through the external schottky diode into the load. after the constant off-time interval, the switch turns on and the cycle repeats. the peak inductor current is controlled by the voltage on the i th /run pin, which is the output of the error amplifier.this amplifier compares the v fb pin to the 1.25v reference. when the load current increases, the fb voltage decreases slightly below the reference. this decrease causes the error amplifier to increase the i th /run voltage until the average inductor current matches the new load current. the main control loop is shut down by pulling the i th /run pin to ground. when the pin is released an external resistor is used to charge the compensation capacitor. when the voltage at the i th /run pin reaches 0.8v, the main control loop is enabled and the error amplifier drives the i th /run pin. soft-start can be implemented by ramping the voltage on the i th /run pin (see applications information sec- tion). low current operation when the load is relatively light, the ltc1701 automati- cally switches to burst mode operation in which the inter- nal pmos switch operates intermittently based on load demand. the main control loop is interrupted when the output voltage reaches the desired regulated value. the hysteretic voltage comparator trips when i th /run is be- low 1.5v, shutting off the switch and reducing the power consumed. the output capacitor and the inductor supply the power to the load until the output voltage drops slightly and the i th /run pin exceeds 1.5v, turning on the switch and the main control loop which starts another cycle. dropout operation in dropout, the internal pmos switch is turned on continu- ously (100% duty cycle) providing low dropout operation with v out at v in . since the ltc1701 does not incorporate an under voltage lockout, care should be taken to shut down the ltc1701 for v in < 2.5v. applicatio s i for atio wu u u the basic ltc1701 application circuit is shown in figure 1. external component selection is driven by the load requirement and begins with the selection of l1. once l1 is chosen, the schottky diode d1 can be selected followed by c in and c out . l selection and operating frequency the operating frequency is fixed by v in , v out and the constant off-time of about 500ns. the complete expres- sion for operating frequency is given by: f o = - + ? ? ? ? ? ? ? ? vv vv t in out in d off 1 although the inductor does not influence the operating frequency, the inductor value has a direct effect on ripple current. the inductor ripple current d i l decreases with higher inductance and increases with higher v in or v out : d= - ? ? ? ? + + ? ? ? ? i vv fl vv vv l in out out d in d
6 ltc1701 applicatio s i for atio wu u u where v d is the output schottky diode forward drop. accepting larger values of d i l allows the use of low inductances, but results in higher output voltage ripple and greater core losses. a reasonable starting point for setting ripple current is d i l = 0.4a. the inductor value also has an effect on low current operation. lower inductor values (higher d i l ) will cause burst mode operation to begin at higher load currents, which can cause a dip in efficiency in the upper range of low current operation. in burst mode operation, lower inductance values will cause the burst frequency to de- crease. inductor core selection once the value for l is selected, the type of inductor must be chosen. basically, there are two kinds of losses in an inductor core and copper losses. core losses are dependent on the peak-to-peak ripple current and core material. however, it is independent of the physical size of the core. by increasing inductance, the peak-to-peak inductor ripple current will decrease, there- fore reducing core loss. unfortunately, increased induc- tance requires more turns of wire and, therefore, copper losses will increase. high efficiency converters generally cannot afford the core loss found in low cost powdered iron cores, forcing the use of more expensive ferrite, molypermalloy or kool m m ? cores. ferrite designs have very low core loss and are preferred at high switching frequencies. ferrite core ma- terial saturates hard, which means that inductance col- lapses abruptly when the peak design current is exceeded. this results in an abrupt increase in inductor ripple current and consequent output voltage ripple. do not allow the core to saturate! molypermalloy (from magnetics, inc.) is a very good, low loss core material for toroids, but it is more expensive than ferrite. a reasonable compromise from the same manu- facturer is kool m m core material. toroids are very space efficient, expecially when you can use several layers of wire. because they generally lack a bobbin, mounting is more difficult. however, surface mount designs that do not increase the height significantly are available catch diode selection the diode d1 shown in figure 1 conducts during the off- time. it is important to adequately specify the diode peak current and average power dissipation so as not to exceed the diode ratings. losses in the catch diode depend on forward drop and switching times. therefore, schottky diodes are a good choice for low drop and fast switching times. since the catch diode carries the load current during the off-time, the average diode current is dependent on the switch duty cycle. at high input voltages, the diode con- ducts most of the time. as v in approaches v out , the diode conducts only a small fraction of the time. the most stressful condition for the diode is when the regulator output is shorted to ground. under short-circuit conditions (v out = 0v), the diode must safely handle i sc(pk) at close to 100% duty cycle. under normal load conditions, the average current con- ducted by the diode is simply: ii vv vv diode avg load avg in out in d () () = - + ? ? ? ? remember to keep lead lengths short and observe proper grounding (see board layout considerations) to avoid ringing and increased dissipation. the forward voltage drop allowed in the diode is calculated from the maximum short-circuit current as: v p i vv v d d sc avg in d in ? ? ? ? ? + ? ? ? ? () where p d is the allowable diode power dissipation and will be determined by efficiency and/or thermal requirements (see efficiency considerations). most ltc1701 circuits will be well served by either an mbr0520l or an mbrm120l. an mbr0520l is a good choice for i out(max) 500ma, as long as the output doesnt need to sustain a continuous short. kool m m is a registered trademark of magnetics, inc.
7 ltc1701 input capacitor (c in ) selection in continuous mode, the input current of the converter is a square wave with a duty cycle of approximately v out / v in . to prevent large voltage transients, a low equivalent series resistance (esr) input capacitor sized for the maxi- mum rms current must be used. the maximum rms capacitor current is given by: ii vvv v rms max out in out in ? - () where the maximum average output current i max equals the peak current (1 amp) minus half the peak-to-peak ripple current, i max = 1 C d i l /2. this formula has a maximum at v in = 2v out , where i rms = i out /2. this simple worst-case is commonly used to design because even significant deviations do not offer much relief. note that capacitor manufacturers ripple current ratings are often based on only 2000 hours life- time. this makes it advisable to further derate the capaci- tor, or choose a capacitor rated at a higher temperature than required. several capacitors may also be paralleled to meet the size or height requirements of the design. an additional 0.1 m f to 1 m f ceramic capacitor is also recom- mended on v in for high frequency decoupling. output capacitor (c out ) selection the selection of c out is driven by the required esr. typically, once the esr requirement is satisfied, the capacitance is adequate for filtering. the output ripple ( d v out ) is determined by: d?d + ? ? ? ? v i esr fc out l out 1 8 where f = operating frequency, c out = output capacitance and d i l = ripple current in the inductor. with d i l = 0.4 i out(max) the output ripple will be less than 100mv with: esr cout < 100m w once the esr requirements for c out have been met, the rms current rating generally far exceeds the i ripple(p-p) requirement. when the capacitance of c out is made too small, the output ripple at low frequencies will be large enough to trip the i th comparator. this causes burst mode operation to be activated when the ltc1701 would normally be in continuous mode operation. the effect can be improved at higher frequencies with lower inductor values. in surface mount applications, multiple capacitors may have to be paralleled to meet the capacitance, esr or rms current handling requirement of the application. alumi- num electrolyte and dry tantulum capacitors are both available in surface mount configurations. in the case of tantalum, it is critical that the capacitors are surge tested for use in switching power supplies. an excellent choice is the avx tps, avx tpsv and kemet t510 series of surface mount tantalums, avalable in case heights ranging from 2mm to 4mm. other capacitor types include nichicon pl series, sanyo poscap and panasonic sp. ceramic capacitors higher value, lower cost ceramic capacitors are now becoming available in smaller case sizes. these are tempt- ing for switching regulator use because of their very low esr. unfortunately, the esr is so low that it can cause loop stability problems. solid tantalum capacitor esr generates a loop zero at 5khz to 50khz that is instrumen- tal in giving acceptable loop phase margin. ceramic ca- pacitors remain capacitive to beyond 300khz and usually resonate with their esl before esr becomes effective. also, ceramic caps are prone to temperature effects which requires the designer to check loop stability over the operating temperature range. for these reasons, most of the input and output capaci- tance should be composed of tantalum capacitors for stability combined with about 0.1 m f to 1 m f of ceramic capacitors for high frequency decoupling. great care must be taken when using only ceramic input and output capaci- tors. the opti-loop compensation allows transient re- sponse to be optimized for all types of output capacitors, including low esr ceramics. setting the output voltage the ltc1701 develops a 1.25v reference voltage between the feedback pin, v fb , and the signal ground as shown in applicatio s i for atio wu u u
8 ltc1701 figure 2. the output voltage is set by a resistive divider according to the following formula: v out = 1.25v(1 + r2/r1) to prevent stray pickup, a capacitor of about 5pf can be added across r1, located close to the ltc1701. unfortu- nately, the load step response is degraded by this capaci- tor. using a good printed circuit board layout eliminates the need for this capacitor. great care should be taken to route the v fb line away from noise sources, such as the inductor or the sw line. applicatio s i for atio wu u u v fb sgnd ltc1701 r2 1% r1 1% 1701 f02 v out c f 5pf transient response the opti-loop compensation allows the transient re- sponse to be optimized for a wide range of loads and output capacitors. the availability of the i th pin not only allows optimization of the control loop behavior but also provides a dc coupled and ac filtered closed-loop re- sponse test point. the dc step, rise time and settling at this test point truly reflects the closed-loop response. assum- ing a predominately second order system, phase margin and/or damping factor can be estimated using the percent- age of overshoot seen at this pin. the bandwidth can also be estimated by examining the rise time at the pin. the i th external components shown in the figure 1 circuit will provide an adequate starting point for most applica- tions. the series r3-c3 filter sets the dominant pole-zero loop compensation. the values can be modified slightly (from 0.5 to 2 times their suggested values) to optimize transient response once the final pc layout is done and the particular output capacitor type and value have been determined. the output capacitors need to be selected because the various types and values determine the loop feedback factor gain and phrase. an output current pulse figure 2. setting the output voltage of 20% to 100% of full-load current having a rise time of 1 m s to 10 m s will produce output voltage and i th pin waveforms that will give a sense of the overall loop stability without breaking the feedback loop. the initial output voltage step may not be within the bandwidth of the feedback loop, so the standard second- order overshoot/dc ratio cannot be used to determine phase margin. the gain of the loop increases with r3 and the bandwidth of the loop increases with decreasing c3. if r3 is increased by the same factor that c3 is decreased, the zero frequency will be kept the same, thereby keeping the phase the same in the most critical frequency range of the feedback loop. in addition, a feed-forward capacitor, c f , can be added to improve the high frequency response, as shown in figure 2. capacitor c f provides phase lead by creating a high frequency zero with r2 which improves the phase margin. the output voltage settling behavior is related to the stability of the closed-loop system and will demonstrate the actual overall supply performance. for a detailed explanation of optimizing the compensation components, including a review of control loop theory, refer to applica- tion note 76. run function the i th /run pin is a dual purpose pin that provides the loop compensation and a means to shut down the ltc1701. soft-start can also be implemented with this pin. soft-start reduces surge currents from v in by gradually increasing the internal peak inductor current. power supply sequenc- ing can also be accomplished using this pin. an external pull-up is required to charge the external capacitor c3 in figure 1. typically, a 1m resistor between v in and i th /run is used. when the voltage on i th /run reaches about 0.8v the ltc1701 begins operating. at this point the error amplifier pulls up the i th /run pin to the normal operating range of 1.25v to 2.25v. soft-start can be implemented by ramping the voltage on i th /run during start-up as shown in figure 3(b). as the voltage on i th /run ramps through its operating range the internal peak current limit is also ramped at a proportional linear rate.
9 ltc1701 applicatio s i for atio wu u u during normal operation the voltage on the i th /run pin will vary from 1.25v to 2.25v depending on the load current. pulling the i th /run pin below 0.8v puts the ltc1701 into a low quiescent current shutdown mode (i q < 1 m a). this pin can be driven directly from logic as shown in figures 3(a) . continuous mode, i gatechg = f ? q p , where q p is the gate charge of the internal mosfet switch. 3) i 2 r losses are predicted from the dc resistances of the mosfet and inductor. in continuous mode the average output current flows through l, but is chopped between the topside internal mosfet and the schottky diode. at low supply voltages where the switch on-resistance is higher and the switch is on for longer periods due to the higher duty cycle, the switch losses will dominate. using a larger inductance helps minimize these switch losses. at high supply voltages, these losses are proportional to the load. i 2 r losses cause the efficiency to drop at high output currents. 4) the schottky diode is a major source of power loss at high currents and gets worse at low output voltages. the diode loss is calculated by multiplying the forward voltage drop times the diode duty cycle multiplied by the load current. other hidden losses such as copper trace and internal battery resistances can account for additional efficiency degradations in portable systems. it is very important to include these system level losses in the design of a system. the internal battery and fuse resistance losses can be minimized by making sure that c in has adequate charge storage and very low esr at the switching fre- quency. other losses including schottky conduction losses during dead-time and inductor core losses generally ac- count for less than 2% total additional loss. thermal considerations the power handling capability of the device at high ambi- ent temperatures will be limited by the maximum rated junction temperature (125 c). it is important to give careful consideration to all sources of thermal resistance from junction to ambient. additional heat sources mounted nearby must also be considered. for surface mount devices, heat sinking is accomplished by using the heat spreading capabilities of the pc board and its copper traces. copper board stiffeners and plated through-holes can also be used to spread the heat gener- ated by power devices. figure 3. i th /run pin interfacing efficiency considerations the percent efficiency of a switching regulator is equal to the output power divided by the input power times 100%. it is often useful to analyze individual losses to determine what is limiting the efficiency and what change would produce the most improvement. percent efficiency can be expressed as: %efficiency = 100% C (l1 + l2 + l3 + ...) where l1, l2, etc. are the individual losses as a percentage of input power. although all dissipative elements in the circuit produce losses, 4 main sources usually account for most of the losses in ltc1701 circuits: 1) ltc1701 v in current, 2) switching losses, 3) i 2 r losses, 4) schottky diode losses. 1) the v in current is the dc supply current given in the electrical characteristics which excludes mosfet driver and control currents. v in current results in a small (< 0.1%) loss that increases with v in , even at no load. 2) the switching current is the sum of the internal mosfet driver and control currents. the mosfet driver current results from switching the gate capacitance of the power mosfet. each time a mosfet gate is switched from low to high to low again, a packet of charge dq moves from v in to ground. the resulting dq/dt is a current out of v in that is typically much larger than the control circuit current. in d1 i th /run c c r c 1701 f03 i th /run c c r c c1 r1 (a) (b)
10 ltc1701 sw v fb v in i th /run gnd ltc1701 d1 l1 r2 r1 c out 1701 f04 v out v in 1 2 3 5 4 c in c c r c r s bold lines indicate high current paths + + figure 4. ltc1701 layout diagram (see board layout checklist) applicatio s i for atio wu u u the following table lists thermal resistance for several different board sizes and copper areas. all measurements were taken in still air on 3/32" fr-4 board with one ounce copper. table 1. measured thermal resistance copper area thermal resistance topside* backside board area q ja 2500mm 2 2500mm 2 2500mm 2 125 c/w 1000mm 2 2500mm 2 2500mm 2 125 c/w 225mm 2 2500mm 2 2500mm 2 130 c/w 100mm 2 2500mm 2 2500mm 2 135 c/w 50mm 2 2500mm 2 2500mm 2 150 c/w *device is mounted on topside. calculating junction temperature in a majority of applications, the ltc1701 does not dissi- pate much heat due to its high efficiency. however, in applications where the switching regulator is running at high duty cycles or the part is in dropout with the switch turned on continuously (dc), some thermal analysis is required. the goal of the thermal analysis is to determine whether the power dissipated by the regulator exceeds the maximum junction temperature. the temperature rise is given by: t rise = p d ? q ja where p d is the power dissipated by the regulator and q ja is the thermal resistance from the junction of the die to the ambient temperature. the junction temperature is given by: t j = t rise + t ambient as an example, consider the case when the ltc1701 is in dropout at an input voltage of 3.3v with a load current of 0.5a. the on resistance of the p-channel switch is approxi- mately 0.30 w . therefore, power dissipated by the part is: p d = i 2 ? r ds(on) = 75mw the sot package junction-to-ambient thermal resistance, q ja , will be in the range of 125 c/w to 150 c/w. therefore, the junction temperature of the regulator operating in a 25 c ambient temperature is approximately: t j = 0.075 ? 150 + 25 = 36 c remembering that the above junction temperature is ob- tained from a r ds(on) at 25 c, we might recalculate the junction temperature based on a higher r ds(on) since it increases with temperature. however, we can safely as- sume that the actual junction temperature will not exceed the absolute maximum junction temperature of 125 c. board layout considerations when laying out the printed circuit board, the following checklist should be used to ensure proper operation of the ltc1701. these items are also illustrated graphically in the layout diagram of figure 4. check the following in your layout: 1. does the capacitor c in connect to the power v in (pin 5) and gnd (pin 2) as close as possible? this capacitor provides the ac current to the internal p-channel mosfet and its driver. 2. is the schottky diode closely connected between the ground (pin 2) and switch output (pin 1)? 3. are the c out , l1 and d1 closely connected? the schottky anode should connect directly to the input capacitor ground. 4. the resistor divider, r1 and r2, must be connected between the (+) plate of c out and a ground line terminated near gnd (pin 2). the feedback signal fb should be routed away from noisy components and traces, such as the sw line (pin 1). 5. keep sensitive components away from the sw pin. the input capacitor c in , the compensation capacitor c c and all the resistors r1, r2, r c and r s should be routed away from the sw trace and the components l1 and d1.
11 ltc1701 typical applicatio s u v in i th /run sw v fb gnd c4 1 f + r3 5.1k r4 1m r1 100k d1 c3 330pf c1 15 f c5 4.7 f + c2 22 f r2 20k l1 4.7 h v in 2.5v to 5.5v v out (1.5v/0.5a) ltc1701 c1: avx taja156m010r c2: avx taja226m006r c4: taiyo yuden lmk212bj105mg c5: taiyo yuden jmk212bj475mg d1: mbrm120l l1: murata lqh3c4r7m24 1701 ta02 all ceramic capacitor 2.5v converter 2mm nominal height 1.5v converter v in i th /run sw v fb gnd c4 1 f + r3 5.1k r5 5.1m r1 20.5k d1 c3 330pf c1 15 f c5 4.7 f + c2 22 f r2 34k r4 1m l1 4.7 h v in 5v v out (3.3v/0.5a) ltc1701 c1: avx taja156m010r c2: avx taja226m006r c4: taiyo yuden lmk212bj105mg c5: taiyo yuden jmk212bj475mg d1: mbrm120l l1: murata lqh3c4r7m24 1701 ta03 on off 5v to 3.3v converter with push-button on/off v in i th /run sw v fb gnd c4 1 f r3 5.1k r4 1m r1 121k d1 c3 180pf c6 33pf c1 10 f c2 10 f r2 121k l1 4.7 h v in 2.5v to 5.5v ltc1701 c1, c2: taiyo yuden jmk316bj106ml c4, c5: taiyo yuden lmk212bj105mg l1: murata lqh3c4r7m24 d1: mbrm120l c5 1 f v out (2.5v/0.5a) 1701 ta01 information furnished by linear technology corporation is believed to be accurate and reliable. however, no responsibility is assumed for its use. linear technology corporation makes no represen- tation that the interconnection of its circuits as described herein will not infringe on existing patent rights.
12 ltc1701 ? linear technology corporation 1999 1701f lt/tp 0800 4k ? printed in usa part number description comments ltc1174/ltc1174-3.3/ high efficiency step-down and inverting dc/dc converter monolithic switching regulator, burst mode operation , ltc1174-5 i out up to 300ma, so-8 ltc1265 1.2a, high efficiency step-down dc/dc converter monolithic, burst mode operation, high efficiency lt1375/lt1376 1.5a, 500khz step-down switching regulator high frequency, small inductor, high efficiency, so-8 ltc1474/ltc1475 low quiescent current high efficiency step-down converter 10 m a i q , 8-pin msop and so packages ltc1622 low input voltage current mode step-down dc/dc controller high frequency, high efficiency, 8-pin msop ltc1627 monolithic synchronous step-down switching regulator so-8, 2.65v v in 10v, i out up to 500ma ltc1707 monolithic synchronous step-down switching regulator so-8, 2.95v v in 10v, v ref output ltc1771 low quiescent current, high efficiency step-down controller 10 m a i q , 8-pin msop and so packages ltc1772 low input voltage current mode step-down dc/dc controller 550khz, 6-pin sot-23, i out up to 5a, 2.2v < v in < 10v ltc1877/ltc1878 high efficiency, monolithic synchronous step-down regulators 10 m a i q , 2.65 v in 10v, msop package up to 600ma related parts single cell li-ion to 3.3v zeta converter typical applicatio u dimensions in inches (millimeters) unless otherwise noted. package descriptio u s5 package 5-lead plastic sot-23 (ltc dwg # 05-08-1633) linear technology corporation 1630 mccarthy blvd., milpitas, ca 95035-7417 (408) 432-1900 l fax: (408) 434-0507 l www.linear-tech.com 0.95 (0.037) ref 1.50 ?1.75 (0.059 ?0.069) 0.35 ?0.55 (0.014 ?0.022) 0.35 ?0.50 (0.014 ?0.020) five places (note 2) s5 sot-23 0599 2.80 ?3.00 (0.110 ?0.118) (note 3) 1.90 (0.074) ref 0.90 ?1.45 (0.035 ?0.057) 0.90 ?1.30 (0.035 ?0.051) 0.00 ?0.15 (0.00 ?0.006) 0.09 ?0.20 (0.004 ?0.008) (note 2) 2.60 ?3.00 (0.102 ?0.118) note: 1. dimensions are in millimeters 2. dimensions are inclusive of plating 3. dimensions are exclusive of mold flash and metal burr 4. mold flash shall not exceed 0.254mm 5. package eiaj reference is sc-74a (eiaj) + + v in i th /run sw v fb gnd c2 22 f + c1 22 f r3 5.1k r4 1m r1 20.5k l2 d1 c3 330pf c4 1 f 5r r2 34k l1 4.7 h c6 4.7 f v in 2.5v to 4.2v v out (3.3v) ltc1701 c1, c2: avx taja226m006r c6: taiyo yuden jmk212bj475mg 1701 ta04 v in i out(max) 2.5v 200ma 3.0v 225ma 3.5v 250ma 4.0v 280ma 4.2v 290ma d1: mbr0520l l1, l2: sumida clq72-4r7 drg no 6333-jps-010

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